That's an interesting old machine - very cool.

One thing though, is that unless I'm missing something, I believe the
two available loop time constants are in minutes, not seconds, or that
it should be in many more (maybe 100X) seconds, if stated that way.
Since the unit can synchronize to a 1 PPS reference, it would make sense
that the loop filtering goes way beyond 1 or 100 seconds.

If this is the case, then there's some typo errors in the manual.

As far as I know, time constant is still T=RC, or megohms X uF =
seconds, in the convenient short form I always remember. So, the long
time constant setting of 2500 megs by 10 uF gives 25,000 seconds - over
400 minutes. Now, I can picture it being defined also by the scaling of
the tuning range used. If you take the input divider 100 k/ 10 k, times
the amplifier gain a little less than 2, that gets it overall into the
100 minutes ballpark. The filter is not an integrator in the pure sense,
but an RC LPF, so the output is bounded to about 20% of the "stored"
tuning voltage from the DAC system.

Regardless of how you estimate, it seems like the times have to be in
minutes, not seconds.

Ed

I'm wondering if anyone has dissected enough common canned RF mixers, to
know how symmetric they are, internal construction-wise, or knows of
available info, especially on specific models.

I have taken some apart over the years, and I believe they generally are
made highly symmetric wrt the LO and RF ports. I typically use them
either way around, for units that have the same band specs on both. But,
the specs typically are with the given port assignments, so there may be
some questions, depending on application.

The particular models in this situation are the WJ M9D, and MCL SRA-1H,
which are high level +20 and +17 dBm, respectively. I have only one M9D,
and two SRA-1H - they're all I have in this class, with a favorable
pinout that I need. I want the IF port to be DC-isolated from ground
(but RF-shorted with C between commons).

Both units appear to be OK for the pinning I need, but with slightly
different arrangement. Having the option of L-R end and phase swapping
(along with IF pinning for best shielding), gives more hookup flexibility.

Since I only have three good candidate mixers, I need to be very careful
to not burn any out, as I'll be driving from an amp capable of over +30
dBm* (with lots of padding and maybe a limiter). Also, the signal input
will be fairly big, up to +6 dBm average, and possibly +20 dBm peak.
This is for that noise source down-converter project I mentioned before.
I'm trying to go as big as possible on signal levels, both to maximize
the output power after conversion and filter loss, and preserve fairly
high crest factor.

The above conditions are about the maximum - in reality, by the time all
the signals are properly padded, the levels will be more realistic. I'm
trying to minimize the padding of course, even looking at using a
diplexer at the IF to absorb the upper image power, to avoid padding the
reflection off the LPF.

*That's at normal 24 V supply. I'm going to try running at 15 V,
unspecified. The maximum Po should be greatly reduced at the lower supply.

Ed

The noise converter project based on the Scientific Atlanta 4647 is
moving along nicely. Still no luck in finding any more info about this
unit, but I did quite a bit of digging in the guts, and figured out
enough to make progress.

The noise generator, to my surprise, is not a typical noise-diode-based
type, but an all-amplifier deal, and apparently the fundamental noise
source is a 75 ohm resistor in conjunction with the input noise of a
2N5179 amplifier front end. The first few stages are broadband, followed
by maybe eight bandpass stages, to craft the power level and shape,
resulting in the 50-90 MHz noise signal, which gets passed to the noise
amplifier box.

The noise amplifier is broadband again, then feeding a CATV type hybrid
power amp for final output, which goes through a ferrite part, which is
either a splitter or directional coupler, for leveling, then on to a
decade step attenuator using Teledyne TO-5 style relays. The leveling
signal from the local detector is sent back to the noise generator box
where it somehow does the gain control. Altogether, a couple dozen or so
transistors are used in the gain stages.

The step attenuator output is sent to the last box, the "C+N amplifier,"
where the external carrier input is attenuated with a step attenuator,
then amplified up and leveled in similar fashion (including another CATV
hybrid PA), then through its own step attenuator, and added to the noise
through a reactive power combiner. So, the noise and carrier signals are
each at least 3 dB bigger than the spec output levels, to accommodate
the adding process.

I added a small board into the noise amp module, with an RF relay to
pass the signal as normal, or route it to the new converter. The maximum
PSD of the noise available there is about -70 dBm/Hz, versus the -73
dBm/Hz at the normal C+N output.

The rest of the action is all built into the 70 MHz oscillator/agc amp
module now. I sacrificed the agc amp function, and utilized the space
for the mixer and LPF, and added yet another CATV type PA in the
oscillator section, for the LO. More on this in the next installment.

Ed

Do you need a 4647?

Lester B Veenstra  K1YCM  MØYCM  W8YCM  6Y6Y
lester@veenstras.com

452 Stable Ln (HC84 RFD USPS Mail)
Keyser WV 26726

GPS: 39.336826 N  78.982287 W (Google)
GPS: 39.33682 N  78.9823741 W (GPSDO)

Telephones:
Home:                     +1-304-289-6057
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-----Original Message-----
From: ed breya [mailto:eb@telight.com]
Sent: Wednesday, May 11, 2022 1:57 AM
To: time-nuts@lists.febo.com
Subject: [time-nuts] Noise down-converter project

The noise converter project based on the Scientific Atlanta 4647 is
moving along nicely. Still no luck in finding any more info about this
unit, but I did quite a bit of digging in the guts, and figured out
enough to make progress.

The noise generator, to my surprise, is not a typical noise-diode-based
type, but an all-amplifier deal, and apparently the fundamental noise
source is a 75 ohm resistor in conjunction with the input noise of a
2N5179 amplifier front end. The first few stages are broadband, followed
by maybe eight bandpass stages, to craft the power level and shape,
resulting in the 50-90 MHz noise signal, which gets passed to the noise
amplifier box.

The noise amplifier is broadband again, then feeding a CATV type hybrid
power amp for final output, which goes through a ferrite part, which is
either a splitter or directional coupler, for leveling, then on to a
decade step attenuator using Teledyne TO-5 style relays. The leveling
signal from the local detector is sent back to the noise generator box
where it somehow does the gain control. Altogether, a couple dozen or so
transistors are used in the gain stages.

The step attenuator output is sent to the last box, the "C+N amplifier,"
where the external carrier input is attenuated with a step attenuator,
then amplified up and leveled in similar fashion (including another CATV
hybrid PA), then through its own step attenuator, and added to the noise
through a reactive power combiner. So, the noise and carrier signals are
each at least 3 dB bigger than the spec output levels, to accommodate
the adding process.

I added a small board into the noise amp module, with an RF relay to
pass the signal as normal, or route it to the new converter. The maximum
PSD of the noise available there is about -70 dBm/Hz, versus the -73
dBm/Hz at the normal C+N output.

The rest of the action is all built into the 70 MHz oscillator/agc amp
module now. I sacrificed the agc amp function, and utilized the space
for the mixer and LPF, and added yet another CATV type PA in the
oscillator section, for the LO. More on this in the next installment.

Ed

time-nuts mailing list -- time-nuts@lists.febo.com -- To unsubscribe send an
email to time-nuts-leave@lists.febo.com
To unsubscribe, go to and follow the instructions there.

Continuing on, the 70 MHz for the LO is tapped off at a leveled low
impedance point, that feeds the normal 70 MHz 0 dBm output on the front
panel. The tap off point is probably around +3 dBm, and I added a higher
R attenuator to get about -10 dBm for the power amp. This CATV amp is
made for 24-30 V operation, but works OK on 15 V, with much less output
power available, and high distortion (obvious on a scope), but still
plenty of gain (35 dB). The output runs about 25 dBm, while the
saturated output power limit is about 28 dBm, which are just about right
for good drive level, but not too much fault power, to avoid mixer
damage if anything goes wrong. The output is already way into
compression, but that's OK. A 6 dB pad connects it to the mixer,
providing nominal drive around 19 dBm, or 22 dBm fault, which is the
mixer's maximum power rating.

That all is what was planned, but what actually shows is that the mixer
looks like a lower Z, well below 50 ohms. I set up the drive with a
built in monitor port that provides a -26 dB view, that showed about
right with a 50 ohm load in place of the mixer, but much lower with the
mixer - it looks like about 15 dBm. It seems to run fine, but is a
little odd. I don't want to push it too hard without more study, so it
is what is is for now.

The maximum noise power comes in at around -70 dBm/Hz from 75 ohms, and
it turns out that a min-loss 75-50 ohm broadband pad is just about right
to knock off 6 dB, putting the R input total power level around +1 dBm,
and peak up to +16 dBm due to crest factor. This is totally safe for the
mixer, and provides good power output. The crest factor will be degraded
somewhat due to running into the LO limit, but only at the highest power
settings. It should be preserved well at lower power.

The chosen mixer is the WJ M9D, which I've discussed previously. Since
this setup is a DSB down-conversion, the conversion loss is less (about
twice as good) than for SSB. I estimate it at around 4 dB, which seems
to agree with my measurements so far. Interestingly, the 50-90 MHz noise
power is not like a typical up-converted baseband signal. Each
"sideband" around the 70 MHz is not redundant to other - they are
independent and uncorrelated (I would think) noise, and simply add
together.

So anyway, ignoring the losses, half of the incident noise power is
converted to the 0 to about 25 MHz range, and the other half goes mostly
to the upper image centered at 140 MHz, and the higher order products.
The IF spectrum viewed on the SA is interesting. The DC-25 MHz portion
is the biggest, and dead-flat in the scale of things. The upper image
looks about 3 dB less, to account for all the rest of the power
contained in the higher products - they are quite large, and go out
quite a way.

That's all for now. Next up will be more mixer and filter stuff.

Ed

Continuing on, the mixer's output looks amazingly good. The filter's,
not so much. I have the IF now going directly to the SA input - no pads,
no filters, no nothing, except some SMB cable/adapter stuff, and about
20 feet of BNC cable. It looks great, letting the SA do the filtering.
The low end is a beautiful down-converted replica of the 50-90 MHz noise
signal.

I can't make high precision measurements here - most are eyeball
estimates from the SA screen, but everything is in the right ballpark,
and makes sense. The amplitude measurements depend on the SA's IF RBW
setting, which is 3 MHz maximum. The measured levels agree well with
different RBW settings. The video BW also affects it some, since extra
filtering is needed sometimes to smooth the curves.

The spec of the 4647 says the effective noise BW is 48.2 MHz. The IF
passes through the -3 dB point near 24 MHz, in close agreement. The
level is very flat (no discernible deviation), to around 20 MHz, where
it just visibly starts to curve into the band edge. The maximum PSD
appears to be around -80 to -83 dBm/Hz, estimated from the displayed
levels at different RBWs.

So, the desired signal is wonderful, if only it didn't include
everything else above. What I need is a very good LPF to get the job
done - the usual problem.

The actual filter I've been using does a good job on the higher
frequencies, but is poor on flatness. It has about 2-3 dB p-p passband
ripple, with periodicity around 5-7 MHz. I've tried various padding
arrangements at both ends, all of which tend to flatten it only a little
bit at best. Looking at it with the TG/SA setup, the character is
intrinsic to filter, and not due to just its reaction to the mixer and
cabling and such.

I hate building filters. Designing them in principle is easy, with all
sorts of available tools online, but actually rounding up the real parts
(and their parasitics) and physical implementation is a PITA. But, I
suppose I'll have to do it eventually for this project. I know how nice
it can be, with the right filter, but for now, I'll have to go with what
I have.

This particular filter is a packaged module type that I've had for a
long time, and used in many experimental setups. In fact, I had to
borrow it from its commitment to another project. Despite its
limitations, it can be very handy, and it is very simple inside, so I'd
like to replicate it for other uses. I plan to open a thread about this
as a separate issue.

In the mean time, it will be for this noise project, and I'll have some
more to report, so next up will be the low frequency/DC aspects.

Ed

Thanks all, for filter info. For reasons that will become evident when I
describe the LF/DC situation, I plan to use an all-passive LC LPF. I
assume I'll be needing a fairly high-order (like 9 or so) Butterworth
type response for good flatness, and enough stop-band rejection for the
higher frequency junk.

The filter that I'm using for now is an old (1970s - 80s?) K&L brand,
marked 4L52-20-0/0-100. I've never found specific info on it over all
the years, but I did just find some data on current products that look
similar:

https://klmicrowave.nyc3.digitaloceanspaces.com/products/attachments/_plk147_1_LLSeries.pdf

The descriptions look familiar enough to get some idea of what it should
do, except the passband ripple is nowhere near these specs (0.05 dB).
The modern part numbering scheme is different, but the "4" and the "20"
seem to jive, for 4-sections, and 20 MHz fc. It looks like K&L describes
the number of sections as the number of choke elements, and that's
what's in this filter. It has 4 chokes and 5 capacitors, so 9th order,
as I understand.

I picked this up in some junk long ago, and it was in bad shape -
someone had opened it up, and the cover was left hanging by a thread -
literally - a single 0-80 screw managed to keep the lid associated with
the rest. I thought the other corners were drilled out, so I just taped
it shut, and since it seemed to work, I started using it for experiments
over the years.

A couple weeks back, I began looking at it closely. I was going to
replace the original SMAs with SMBs for this project, so figured I'd
pull the guts out so I could try to ID the part values. The caps were
easy, just regular mica types with markings. The chokes are small
ferrite toroids, apparently identical cores. I counted the turns, and
found 20 on the outer pair, and 22 on the inner. I didn't want to risk
damage by removing any parts (the chokes are silicone-gooped to the
shielding), but I did manage to get ballpark in-circuit values using the
HP4276A LCZ meter at 20 kHz, so the error from the caps isn't too bad.

I also had a bit of good luck in finding that a screw was loose - one
that anchors the assembly to the floor of the machined Al box, and is
critical for grounding the circuit board. I had assumed this was not all
that great of a filter, barely keeping the stop-band 40 dB down, or that
maybe it would get better if the lid was properly attached. I also found
that there was enough intact thread left in the corner holes, that
digging up some 0-80 screws of just the right lengths fixed the lid
mounting.

So, I got my values, got my SMBs, and the filter is rebuilt almost like
new. It still leaks some at the higher frequencies, but it's now better
than 70 dB down, which is on par with the modern spec, which only shows
to that level. It is a fairly sharp cutoff filter, dropping about 80 dB
at 50 MHz

Here's the parts info:

Caps (pF labeled, unknown tolerance) 68, 150, 150, 150, 68
Chokes (uH +/- 20% possible measurement error) 2.25, 2.43, 2.5, 2.28
Choke ratio inner/outer 1.21 according to turns count

So, anyway, I know it's symmetric, supposedly 50 ohms, and 20 MHz fc.
Since then, I've been looking at filter design tools, trying to match
what's in there to any kind of "standard" filter response, and tweaking
fc and impedance too. So far, I've found nothing that's close.

The actual amplitude response looks very much like the Chebyshev example
that Gerhard posted, and the datasheet says that's what this product
line is, so it's probably in there somewhere. It's just that the part
values don't make sense.

Ed

I managed to build a filter, using the values for a 9th order
Butterworth, 50 ohms, 25 MHz fc. The caps were fairly straightforward to
get nearly right on in values, with one or two (paralleled) selected
micas for each spot. The chokes were tricky. I decided to use IF-can
style adjustable ones, since I managed to scrounge up a few that were
close enough. The whole thing is built on special double ground plane
board (with 0.15" via grid shorting the sides together) stock, which
took a lot of hand crafting to mount the cans and lay everything out right.

I checked it with the TG, and it looks like a filter, kind of as
expected. After much tweaking of the chokes, I got it to look fairly
good, but it's all open-loop, part-wise - the chokes are set for
appearance of the response, not necessarily right values. So, it's some
kind of LPF, but that's about all I can say. The chokes are the weak
link, since they're hard to measure accurately.

I put the filter into the noise project, and the result looks pretty
good. Measuring the actual noise output on the SA, and zooming in, I
found it was flat to less than half a dB p-p, looking at 1 dB/div. Not
bad considering my eyeball-controlled adjustment using 10 dB/div and the
TG beforehand. This flatness is the net effect of the noise output
itself, and the filter, and a little bit the SA, so pretty decent. The
fc is around 22 MHz at the "best appearance" setting, and the Z-match
seems OK. There's no pad at the filter input, and about 3 dB at the
output, then that same 20 feet of cable to the SA. The high frequency
rejection looks pretty good too, with the 70 MHz and 140 MHz (the worst
offenders) below -85 dBm. This can be improved with more grounding
enhancement, and possibly adding shielding - it's kind of open
construction now, just on the board. The chokes are fairly well
contained and shielded in the cans, but the caps are exposed.

Anyway, for this purpose, it's way better than the original filter,
which can now be returned to its other project. I'm fairly happy with it
so far, but expect it to be one of those never ending projects - always
room for improvement.

Ed

Thanks Mike, for info on LCR alternatives. It's good to know of others
out there, if needed. I have an HP4276A and HP4271A. The 4276A is the
main workhorse for all part checking, since it has a wide range of LCZ,
although limited frequency coverage (100 Hz - 20 kHz). The 4271A is 1
MHz only, and good for smaller and RF parts, but very limited upper LCR
ranges. I think it works, so I can use it if needed, but would have to
check it out and build an official lead set for it. I recall working on
it a few years ago to fix some flakiness in the controls, so not 100%
sure of its present condition.

The main difficulty I've found in measuring small chokes is more of
probing/connection problem rather than instrument limitation. For most
things, I use a ground reference converter that I built for the 4276A
many years ago. It allows ground-referenced measurements, so the DUT
doesn't have to float inside the measuring bridge. The four-wire
arrangement is extended (in modified form) all the way to a small
alligator clip ground, and a probe tip, for DUT connection, so there is
some residual L in the clip and the probe tip, which causes some
variable error, especially in attaching to very small parts and leads.
When you add in the variable contact resistance too, it gets worse.
Imagine holding a small RF can (about a 1/2 inch cube) between your
fingers, with a little clip sort of hanging from one lead, and pressing
the end of the probe tip against the other lead. All the while, there's
the variable contact forces, and effects from the relative positions of
all the pieces and fingers, and the stray C from the coil to the can to
the fingers. I have pretty good dexterity, and have managed to make
these measurements holding all this stuff in one hand, while tweaking
the tuning slug with the other.

I had planned on making other accessories like another clip lead to go
in place of the probe tip, but not yet built. I also have the official
Kelvin-style lead set that came with the unit, so that's an option that
would provide much better accuracy and consistency, but the clips are
fairly large and hard to fit in tight situations, and the DUT must
float. Anyway, I can make all sorts of improvements in holding parts and
hookup, but usually I just clip and poke and try to get close enough -
especially when I have to check a lot of parts, quickly.

The other problem is that the 4276A is near its limit for getting
measurements below 1 uH, with only two digits left for nH. The 4271A
would be much better for this, with 1 nH vs 10 nH resolution.

If I get in a situation where I need to do a lot of this (if I should
get filter madness, for instance), then I'll have to improve the tools
and methods, but I'm OK for now, having slogged through it this time.

Ed

Now that I have the "official" filter in place, I can wrap up the LF/DC
issues. This is the other extreme, so no SA here, just time domain view
with a Tek 7A22 vertical, which gets down to 10 uV/div, and has settable
BW steps from 100 Hz to 1 MHz. For very low f and DC, I use a HP3456A.
There are some limits, especially in the 7A22, which is a little flaky,
but mostly puts on a good show. In either instrument, there may be
errors caused by the large HF part of the noise up to the 25 MHz or so,
way beyond what they're trying to see.

One thing that immediately showed up is the mixer DC offset (about -1.2
mV) due mostly to imperfections in the mixer, distortion in the LO, and
LO leakage into where it doesn't belong. I built a photo-voltaic
circuit to generate a current to cancel it out, but had to wait until
other issues were settled before final design adjustments.

Why a PV generator? This relates to the fundamental design plan. You may
recall that in the earlier talk on the mixer, I wanted to be able to
have galvanic isolation of the IF port, in order to eliminate or reduce
ground loop interference. Indeed, I found out right away that this was
the way to go. On the 7A22, I could see several mV of line-related junk,
and figured it was time to lift the IF off ground. For the RF
experimenting, I had the IF chassis-grounded, but had all the provisions
in place to float the whole works, from the IF port all the way to the
front panel BNC. I chose to overdo the capacitance from the IF common to
earth, with two 100 nF caps. The common-mode chassis noise disappeared,
as expected.

But, all this forces various compromises between the requirements.
First, there's not much point in making a thing that can go essentially
all the way down to DC, and possibly at very tiny signal levels
(depending on BW and noise power level), if you can't convey the signal
to an external piece of gear or experiment without ground loop
interference. So, this isolation is necessary - it raises the
common-mode impedance of the source so that the (hopefully) small
inter-chassis voltages can't push much current between equipment.

But, this is all frequency dependent too. If the ground loop
interference has higher frequency content (like in something with a SMPS
that's not very clean), the caps isolating the floating section present
much lower CM impedance, allowing more current. For this, you'd want
minimal CM capacitance.

But, minimal CM capacitance is minimally effective in shorting out the
LO and RF at the mixer - whatever leaks through due to the limited
isolation of the mixer becomes CM and additional IF signal at the IF
port. For this, you'd want as high a CM capacitance as possible, or
solid ground (which is the non-isolated form).

So, it all boils down to making appropriate trade-offs in that CM
capacitance. As mentioned earlier, I started with 200 nF, which was
sufficient for line/harmonic interference rejection, and was a good RF
short at the mixer. Next, I tried a lower extreme of 2 nF total, which
would have been great for medium frequency rejection, but alas, not a
good enough short for the LO and RF, indicated by increasing power at
the output, and increasing DC offset - it nearly doubled it.

The present compromise is about 9 nF total (the previous 2 nF plus three
2200 pF tacked on). This seems to be pretty good, with reasonably small
(maybe -90 dBm) LO showing, and only slightly higher offset compared to
the 200 nF version. I think when all's said and done, I'll end up with
about a 10-20 nF compromise value.

There's also some CM choking involved. The most important one isolates
the LO and RF CM right at the IF port, formed with three loops (about 10
uH) through a ferrite toroid of the SMB pigtail cable the goes to the
filter. A second one will be included on the output cable to the front
panel, to help at the medium to high frequencies.

I edited the box's board ground plane to form the isolated section that
carries the filter, padding, associated interconnects, and PV generator
circuits. Since this all floats, the PV method is used, and no power
supplies or chassis ground returns (which would spoil it) are needed.
The generator is two paralleled 4N37 opto-isolators operating in PV
mode, with variable LED drive for setting the offset current.

The concept of "floating" is somewhat arbitrary. In reality, the whole
output could float to any applied voltage until something breaks down,
but I decided it was safest to just hard-clamp it to chassis ground with
Si rectifiers (1N5401). Unfortunately, their zero-bias capacitance adds
to the total CM capacitance, while they can't help with any RF shorting
at the mixer - they're too big to fit near there, and are too far
removed from the action by distance and the CM choke.

Next up will be more details. It's getting close to the end. I can tell
that it's near time to wrap up or quit this project, because the
connectors are starting to wear out from all the puts and takes of the
box into the instrument - I'd say it's well over a hundred times already.

Ed

I've been working on final design cleanup, mainly in the RF. I found
quite a bit of spurious LO harmonic content up to almost 2 GHz, with
some quite strong (-75 dBm). It was time to clean up the experimental
wiring layout, so I simplified the cabling and consolidated the RF stuff
onto the LPF board. This improved things a bit, but some spurs were
still pretty big. I presumed most of it was going right through or
around the LPF, and some due to common-mode and cavity resonances inside
the box, which can have many modes.

I added a small LPF about 300 MHz (10 pF/50 nH/10 pF), inside its own
tiny shield box, forming the last bastion of filtering, right at the
inlet of the pigtail cable that goes to the isolated SMA bulkhead
fitting, and including another CM choke (only 1 pass of cable). This
filter is high enough up (over ten times the fc of the main LPF) that
they shouldn't interact very much - they are isolated only by the 3 dB
pad in between.

All along, I've wondered what to do about the reflected power from the
main LPF, that mostly has to go back to the mixer. They are separated by
maybe 300 pSec of cable, which could be in the range for resonances at
the upper end. But, various experiments during development, including
padding the LPF input, and even making a diplexer with a 50 MHz HPF to
take the HF content into a terminator, showed no difference in the noise
output flatness, although the spurious levels likely would have changed
a little - some up, some down. So, I decided to keep it simple and just
let 'er rip, with nothing extra at the LPF input.

Things are now at levels where the fine (and subtle) details show,
mostly cable dress, and grounding. I'll probably be adding bits of
shielding here and there, and maybe fooling with some RF absorbing foam
to see if any box resonances are a problem.

Speaking of subtle effects, here's something interesting. The little
shield box for the 300 MHz LPF is a type with a fold-down lid, on a
hinge formed by thinning the sheet steel. It's only good for a few open
and close operations before the hinge breaks apart, so I kept it open
while building and testing the filter. It looked great, and the time
came to close everything up and look at the spurs again. I closed the
lid, and bent the retainer tangs a little, for good closure. Virtually
all the higher frequency spurs got a few dB worse. So, was it that the
lid isn't really grounded thoroughly, and acting as an antenna to bypass
the filter, or did it affect the choke Q or part values enough, or is it
that I also changed the cable dress a bit while putting it all back
together? I'll have to figure it out.

Anyway, it's looking pretty good right now. With everything closed up,
including the box lids, as it would be when completed, all the spurs
show around -90 dBm or less. There were maybe two dozen noticeable spurs
identified earlier. Some are now in the noise floor (around -105 dBm,
but some remain, sticking out. I think most will disappear if I figure
out that 300 MHz filter box lid, which would leave the 70 MHz as the
main offender. This isn't surprising, since it's the biggest signal of
all, and it's not filtered all that much - it's too close to the main
LPF fc, and below the 300 MHz LPF. I should be able to knock it down
enough with detail work mentioned above, and I'm also pondering ways to
make a 70 MHz trap, if it won't go away. I have a couple of 70 MHz
crystals, so I could try this kind fairly easily. Does anyone have any
handy design info for crystal notch filters in this frequency range? For
an LC trap, it looks like a single L and C would be enough to get the
job done, without interacting too much with the other filters.

Ed

Continuing with experiments and spur measurements, I found that closing
the lid on the little filter box does seem to reduce the LPF's
effectiveness at the higher frequencies, but leaving it open reduces
effectiveness at the lower. I can sculpt it to a taller structure if
necessary, which would give more clearance to the choke, and allow for
some microwave absorbing material. Yes Askild, I did manage to squeeze a
little strip into the existing can, but only near the output end -
there's no room above the parts, especially the choke, so pinching
anything there would likely spoil the whole thing. For now, I have the
lid closed, and the absorber strip at the output area. The hinge of
course has already broken, but I just tack some solder gobs over it as
needed.

What I discovered next, however, may mean I won't have to improve this
box anyway. Doing a spur review, I found that the remaining significant
ones were the 70 MHz, and all but one of the ones between 1260 and 1820
MHz. Above and below, everything else was in the noise floor. I had
gradually worked the 70 MHz down some with shielding and such, but a
little remained even all closed up.

I started thinking again about possible resonance of the cable from the
mixer to the LPF. The length is in the right ballpark to aggravate the
problem spur range. The only reason for the length was to get the
desired three turns on the CM choke, so one option was to give up most
of the choke value and go one turn, with a short, straight connect to
the LPF, which would force any resonances way upward (but maybe they'd
just show up elsewhere, if there's not enough loss at the higher
frequencies). The other option was to revisit padding the input of the
FPF, or the diplexer again.

After thinking back over previous experiments with these, I recalled
that I was really only looking for noise flatness then, and hadn't even
gotten to detailed spur measurements. I also recalled that the original
diplexer setup did interact some at the top of the noise band - I chose
a 5th order 50 MHz .05 dB Chebyshev response, but the real parts made it
something a little different. I realize now that I should have just
disconnected it and left it in place - unfortunately, I took it all out
during my last cleanup and consolidation round.

So, not wanting to change too much around, and only guessing about the
cable situation, I figured on trying something simple and quick to
diplex out the upper stuff, to suppress possible resonances. I chose
somewhat arbitrarily a 3rd order Chebyshev around 140 MHz, which is
where the upper image lies, and the choke is about 50 nH, and I just
happened to have another of the same part I had measured around this
value and used in the 300 MHz LPF. So, in went a 22 pF/50 nH/22 pF/51 R
HPF, and out went the reflections.  All the bad spurs are in the noise
floor in a broadband view, but can still be found with narrow band spot
checks, around -95 to -100 dBm. The net reduction from previous "feels"
like maybe 6 dB, which I think corresponds with cutting VSWR in half.
Interestingly, the 70 MHz is now virtually gone too.

So, it looks like my first instinct to have a diplexer was right, but I
didn't study it deeply enough, and my assumption about its effect on the
spurs being small was wrong. Now that I can see some results, I can set
the HPF a little lower to at least terminate the entire upper image
(about 115-165 MHz), but not so low that it interferes the LPF response.
Another interesting thing is although the upper image is the second
biggest after the desired output signal, it has never shown above the
noise floor since I installed my new LPF to replace the commercial one.

Ed

Yes, that transformer sure looks burned out. It's hard to tell how big
it is from the pictures, but my impression is that it looks kind of
skimpy to run a FRK Rb plus whatever else is going on like a GPS RX and
uP system, and maybe battery charging too. You can easily estimate the
VA rating by measuring the dimensions and comparing to standard
transformer frame sizes. Generally, the VA rating should be at least
twice the total raw (not the regulated output values) DC power produced,
with conventional rectification and filtering. This can be exceeded for
a while, say during warmup of the Rb, as long as it goes back to normal
in a reasonable amount of time. It's mostly about temperature rise - if
you have good cooling, you can get more out of it.

Transformers are pretty tough, so having one burn out in normal service
calls for some investigation of why it happened, before you risk taking
out a replacement too.

Regardless of the VA rating that should be used, you're probably stuck
with using the same size and style as the original, just to fit it
mechanically. If it's plenty big enough VA-wise, then all's well. If
it's marginal, you can at least add enhanced protection to avoid another
burnout.

Regarding DC supply voltages, the main one will be something around 24 V
for the Rb. I would guess that the DC-DC converter on the supply board
makes +5 V (or 3.3 or whatever) for the brain and GPS RX, and the 78M12
makes +12 V for the analog, and that there are no negative supplies -
unless there's more to the supply system that's not shown. Since
external 24 VDC can supposedly run the whole thing, I don't think you'd
have to worry about making any of the voltages from the AC transformer
except for the 24 V, even though it appears to have a multi-tapped
winding. I didn't see anything in the OP about whether the thing works
with just external DC, so this should be confirmed.

There's a lot more circuitry on the board than seems necessary just for
power, so it may be worthwhile to reverse engineer it a bit - especially
the four big transistors and U3 and U4, which looks like two identical
functions of some sort. Maybe extra voltage regulation, or maybe 1 PPS
amplifiers?

Once you do figure everything out and get a fresh transformer, note that
the original is banded to reduce magnetic emission. It appears to have
both the copper strip around the bobbin zone, and the steel (or
sometimes mu-metal) band around the core, but not the third thing
commonly done, which is insulating the core mounting. It will function
without these, but may interfere with the Rb unit, especially if it's
nearby. You won't find these features in run of the mill OEM replacement
transformers, so you'd have to specify them, or add them yourself. If
you get a transformer with same dimensions as original, you can
transplant these pieces from the old one.

Ed

This may give some idea of how fast things can happen when the OCXO is
subject to drafts. I have this dual GPSDO box that usually is open for
experimenting, and have a setup comparing one of the 10 MHz outs to my
portable Rb reference. The 10 GHz multiplied output from the Rb is
indicated on a microwave counter, using the GPSDO as reference. This
gives 1 mHz resolution on the 10 Mhz signals at the 1 Hz counter
resolution limit. It normally reads 10 GHz "exact" +/- 1 Hz when things
are stable, or up to maybe up to 2 Hz when garage ambient is changing. I
just turn the counter on whenever I'm in the mood to take a look.

The upper GPSDO board is exposed, so I can just put a finger on the case
of the small (about 1" x 1.5") OCXO for a few seconds. Almost
immediately, the counter shows several Hz change, which gradually
recovers, with some over- and under-shoot. During all this, the OCXO is
changing, and the GPSDO is trying to fix it.

Having a bigger OCXO with more thermal mass and insulation, and having
more protection from fast ambient changes can help a lot. As others have
said, you don't want to overdo it - the oven heating system must be kept
working under all conditions, but it's OK to make it not have to work
too hard.

An extreme example of a bad thermal situation is in the beloved HP8566.
I have often lamented about the poor placement of its internal OCXO,
which is right in the main air plenum that feeds the fan cooling air to
the whole instrument. The OCXO is subject immediately to any change in
ambient, and its heater has to work very hard. I'm convinced that this
is the cause of most OCXO failures in the 8566. I've had to refurbish a
number of these. The typical failure I've encountered is that the foam
insulation deteriorates from the high heat flux needed, and the
chemicals from the foam cause the oven setpoint adjustment pot wiper
contact to fail. An easy way to spot this problem is to gently shake the
OCXO - if you can hear and feel the guts clunking around inside, then
it's due for repair.

At an opposite extreme, in my "Z3801A in a HP5065A carcass" project, I
substantially isolate the OCXO from ambient. It's already a double-oven
style, and I further enclosed it in a mu-metal box (made from a CRT
shield). The OCXO is suspended on rubber vibration mounts, inside the
box, and has a thin (~1/4") layer of non-woven fiber insulation on all
sides between it and the box. The insulation has very little R-value,
but suppresses turbulence and convection flow inside. The Z3801A guts
are arranged specially to fit and occupy about two thirds of the cabinet
volume, and this section is largely sealed off from the outside and from
the right side battery compartment. A small fan runs at very low speed
to gently circulate the air inside the compartment, and the plentiful
amount of cabinet skin easily dissipates the total power. The same type
of insulation is also placed under and atop the main board in the
DAC/EFC circuit area, to slow down thermal changes there. The EFC's SMB
connector set will also be shrouded with an insulating tube, to reduce
thermal voltage. I even changed the nearest board mounting post to
plastic, to reduce effects of thermal conduction and ground current in
the vicinity.

All of this does not protect from ambient, but only the rate of change.
It's more or less a constant temperature rise type deal, assuming
constant power dissipation when everything's stable - and not too much
wind or draftiness on the whole cabinet.

Ed

Erik, I'd really recommend that you use a real, "solid" ground reference
on the instrumentation side, with +/- large (12-20 V) supplies, as
others have suggested.

Your most recent setup diagram indicates that you're relying on the
"differential" input of the audio PC card etc analyzer to allow for the
"floating" common of the analysis circuit. Do you know what the
common-mode rejection characteristics are? A true differential input
would have two coax lines entering a symmetric differential to
single-ended conversion stage at the front end. I doubt that the PC card
actually has this, but maybe some form of DC/LF isolation from the local
input common to chassis ground.

The PC likely has lots of SMPS noise in common-mode form, which probably
can be ignored for audio (the SMPS frequencies are almost always quite
far above audio). As long as the interference signals aren't too big to
upset the LNA operation by say, rectification in various junctions
(especially the front end), it should be OK. You will also have in-band
line frequency and harmonics present in the common-mode signal, but
these should be easier to deal with by virtue of whatever LF CMRR the
sound card does have at lower frequencies.

Now consider the analysis circuit environment, where you have apparently
zero intentional bypassing capacitance from the floating measurement
common to chassis/earth ground. Here, the only bypass caps effectively
are C1 at the REF buffer's input (which will only aggravate the
situation), and the small capacitance between the ports of the mixers. I
believe you have some bypassing at points in the other portion of the
circuit - the PLL for the reference - but I don't know what that looks
like now. So, just looking at this section, I'd say you need some
serious bypassing to ground, for the RF signals from the mixers, and the
common-mode signals in and out of the audio analyzer, DUT, and REF.

I recall there were some recent discussions about rail-splitting and
such, but I didn't look closely. I thought surely someone would have
mentioned the simple way to rail-split with an opamp, into a large
capacitive load, but maybe not.

Without resorting to a more desirable ground-referenced, +/- supply
scenario, you can add significant bypass capacitance from the signal
common to ground, with slight change to the buffer circuit.

1. Add a resistor between the opamp's output and the load, which is
   signal common. The current demand appears small, so maybe around a
   couple to few hundred ohms should do.

2. Add a resistor in series with the (sense line) inverting input. This
   can be in the many k ohms range, depending the opamp's bias current.

3. Add a small capacitor between the opamp's output and inverting input
   to stabilize it.

4. Add the bypass cap.

This setup just isolates the opamp from the capacitive load, with the
LF/DC regulated by the opamp, and the HF shunted by the bypass cap.

I'm guessing that once you get good bypassing here, the LNA will work
much better, and you should see the difference with the lower noise
opamp. The reason is that any opamp has limited CMRR, so improving the
bypassing makes the "CM" part smaller. This is also another reason to
operate opamp inputs at or near ground. Actually, the best CM
improvement can be provided by running in inverting mode, so both inputs
are always at ground. Non-inverting modes require the inputs to move,
depending on the signal. In your LNA, the CM input signal range is not
too bad, due to the high gain. The trick is to keep the overall CM - the
operating common level wrt ground and the power supplies - constant and
noise-free.

Regarding microphonics, since you mentioned tapping the housing, it
sounds like you have "canned it up," which is a good thing. Assuming the
REF and DUT are external, so not involved, the audible is coming from
the analysis circuit only, right? That's not too surprising since it's a
high gain system. It could be related to individual component
microphonics, but I'd guess it's an RF effect. The whole thing is awash
in the 2f signal and harmonics from the mixer, and to a lesser extent
the DUT frequency signal that leaks through, so mechanical dimension
changes or movements in the can, board, wiring etc, can change the EM
pattern inside, giving tiny, noticeable phase shifts - after all, that's
what it's for.

Ed

I forgot to mention that you should also consider possible effects from
the RF present, on the LNA. This can be more significant than SMPS
frequencies getting where they don't belong, especially since the RF is
intentionally right at the opamp's input. Your LPF only reduces, and
does not eliminate, the 2F and harmonics, so there can be significant RF
present on the LNA circuit.

A simplistic view is that the RF is far beyond the opamp's GBW or closed
loop gain and should have no response, but it's not at all beyond
upsetting or altering the operation. This can result in extra DC offsets
and noise due to RF rectification in the input circuits, which only
remain "linear" at frequencies where the output and feedback can keep up
with the input.

This can be fixed if necessary, by adding extra RF filtering,
particularly some built to low-pass at a higher cutoff frequency well
above the analysis frequency, and well below the expected f and 2f.

For instance, in your circuit it looks like L1 is 1 mH, with 100 nF
caps, which ideally cuts off quite low. However, 1 mH is a pretty big
choke, and will tend to have a lot of inter-winding capacitance (and
high resistance - don't forget to include it in noise), making it less
effective at the higher frequencies. Adding an LC section in front of
it, but set up for something in the MHz region, will give much greater
rejection of the f and 2f, due to having more appropriate smaller L and C.

Anyway, if it works fine as is, then no problem, but it's something to
be aware of if you get strange effects down the road.

Ed

Ed,
Thanks for the many good advice.
I've tried to incorporate as much as possible, updated schematic can be
found here: http://athome.kaashoek.com/time-nuts/PNA/SSPNA.JPG

For audio into the PC I'm using a professional balanced microphone to
USB input with a noise level of -130dBc/Hz and no spurs.
Using a 7805 and 1500uF capacitors I tried to create a solid reference
instead of the buffer op amp output and that did make a difference.
Further the input of the first opamp has been change to have identical
resistors at both the + and - input to reduce common mode signals.
None of the capacitors (5.6pF) I tried to reduce high frequency gain
improved the results. Most of the time it got worse.
Removal of a ceramic capacitors eliminated the microphony.

I've added a switch to select between 0dB and 20dB gain so I can
calibrate the level by offsetting the DUT frequency while keeping the
drive to the mixer.
To calibrate the effective noise BW of the FFT I create a test signal
combining a -70dBm 10.001MHz signal with a -90dBm/Hz noise signal.
The 20dB power ratio was confirmed using a calibrated spectrum analyzer.
The FFT length and sample rate at the PC where then changed till the PC
FFT showed the same power ratio.

Noise level at 1kHz is now -150dBc/Hz and -155dBc/Hz at 10kHz which is
the spec of the DOCXO used so no need to go any lower.
Also the 50Hz spurs and its harmonics are greatly reduced.

I'll have to invest in better coax cables as the current cable seem to
leak RF.
Erik.

Erik, that looks pretty nice, with good stout bypassing. I'd recommend
adding some ceramic caps bypassing for RF, just in case. Also, I believe
the standard three-terminal regulators have no limit on capacitive
loading, so you probably don't need R1. The standard 7805 can put out a
lot of current (>1A), so I'd suggest using the 78L05 low power version
(100 mA or so). That will make it a little safer, fault-wise. Especially
when building and experimenting, it's good to have some limiting
appropriate to the circuit loads. If you should say, inadvertently short
the circuit common to ground, a way bigger than needed regulator will
supply lots of current, and possibly crash the +12V if it can't keep up.
Finally, whenever these regulators have large output capacitors, it's
good to have a reverse protection diode from output to input, in case
the input voltage is suddenly dumped - another thing that can easily happen.

So, with good DC and LF regulation, and plenty of audio and RF
bypassing, the common line should be very solid. I don't think the very
low frequency regulator noise or DC drift will cause any grief, since
the opamp's CMRR should take care of it. I looked up the OP-27, and it
shows over 120 dB CMRR up to a few kHz, so the CM to input signal
conversion is less than 1 uV/V. You should check all the part specs to
be sure if it will be OK. The PSRR is much weaker, so the cleanliness of
the +12V supply should be considered too.

I see that you've changed the LNA circuit (to inverting) and feed to the
audio (ground-referenced - nice). Also, it appears that 10X voltage gain
is sufficient, instead of the 100X previous. This should greatly reduce
the grief. One thing that's different about the inverting mode versus
the previous non-inverting, is that now you have a definite DC/LF
termination/load resistance (the 100 ohms into the summing node) on the
mixer IF port, while the non-inverting mode can provide very high input
R, so you can have more flexibility on termination R choice, from "open"
on down to whatever you want. I don't know what the ideal is, but as in
previous discussions, it doesn't need to be 50 ohms as in RF work, but
can be much higher for the audio, to get a bigger voltage signal. For
inverting mode, the input R must be all or part of the termination R,
and "open" is not an option.

I'm wondering what the PC USB audio box is. Could you please let us know
make, model, etc? The "balanced microphone" description sounds like it's
600 ohm input R.

Ed

Erik, it sounds like you have it nearly finished and working - congrats.
You should be able to greatly reduce those pesky line-interference
spurs. The first weak link may be the audio signal path, so it's time to
let the audio box help, if you haven't already optimized it. In the last
design round, you now have the audio box driven from a coax line with
its shield connected to your local instrumentation ground, which is
earth ground there, and the minus end of the 12V supply, right?

How is the other end hooked up to the audio box? I looked at some info
on the unit, but it doesn't say anything about hookup details - a
"normal" user (not us) would probably just plug in a mic and go. If it
is a balanced mic connection, then it is at least some form of
differential input, and you had been using it this way before, with the
audio common (low side) connected to your floating signal common. That's
the way you should connect the audio box still, even though the cable is
now earth ground referenced at the the instrument box. If it's still set
up this way anyway, then OK. If you have connected the cable shield to
the ground at the PC or USB audio box, try removing it. The audio box
input should have some amount of LF CMR, which should help in this part.
You can experiment with different arrangements and see what works best.
The next thing is to try adding common-mode choking with various methods

• I'll explain later.

From the very beginning, I think, you have considered the benefits of
galvanic isolation of the DBM/PD mixer ports, which is good.
Unfortunately, powering up the OCXOs and getting their signals connected
leads to some inevitable ground loops anyway, so you can't always take
full advantage of built-in isolation, but at least you have it. Try to
use it when you can.

The rest of the line and low frequency ground loop issues are hard to
predict, depending on the overall powering and operational setups, but
some general methods can be applied.

Now back to the audio output path, the first thing is to see if there's
any point to improving its CMR - if the interference is getting in
elsewhere, it doesn't matter (yet) how good it is. You can experiment
with the circuit hookup and operation. For instance, in your latest
post, the first chart shows the operating noise floor and line spurs
present "without the DUT connected." This can describe a range of
possibilities, from the RF connector being unplugged, to the DUT being
totally removed from the experiment, power-wise, ground-wise, and so on.
In your setup, you have a number of items hooked up, each contributing
to the ground loop situation.

You can apply the process of elimination to gradually improve it. The
first and easiest step is to shut everything down except the PC and
audio analyzer setup, and assess the spurs. If there's little change,
then the interference is what it is, due to the environment of the
setup. If the spurs go away, then your analyzer circuit is amplifying
them from the signal chain, or from the power supplies. Then you would
short the input of the LNA, power it up and look at the spurs again, and
so on.

This narrows things down as you go, with various tests to see what
things make the biggest difference. If the spurs are relatively
unchanged with power down, then you would unhook things until you're
left with just the analyzer circuit hanging from the cable to the audio
box - you systematically delete connections to power supplies, grounds,
signals etc, until it shows clean. Then you can add back and think about
the possible ways the interference gets in.

Once you figure out the major interference sources, you'll know what and
where to make improvements. Keep in mind that within your own project
circuit, you have lots of options, since you're building it, while on
most of the other stuff, you'll likely be limited to external fixes like
CM chokes (prayer beads) and grounding tricks.

Next time I'll talk about some of the methods in detail.

Ed

I've been scrounging around in the OCXO department and found two of
these that I likely pulled from something junked out long ago. I found
the MTI website, current catalog, model line etc, but of course no
mention of this specific version, or anything close, numerically. I
can't make any sense of the sub-model numbers, so assume this was a
custom version for somebody, or long obsolete. Does anyone know of this
particular model version, or if there's a "system" to the sub-model
numbering scheme?

Frequency 10 MHz
Model 250-0709-B
Part # 010209-0709

I believe the date codes are 1997 and 1998

The notes I made on them say they're +12V powered, which I must have
figured out back when I pulled them.

One thing different about these versus the current ones pictured is they
have no screw-plug port to access mechanical/coarse tuning - they're
solid all around. It might be that the "-B" suffix deletes this access.
I suppose that this could be a good or a bad thing, depending on use.
Maybe it's very stable, long term, or maybe it has a very wide EFC
tuning range.

I'm hoping it's an SC-cut, sine out, for lowest available phase noise
and utility for my needs. Worst case would be AT, HCMOS, poor phase
noise and excessive tuning range, so of course, that's most likely.

If no info is available, I can fire it up and check the output type easy
enough. Also, as I recall, you can tell the cut by the frequency
behavior during warm up, but I forget which does which. Any suggestions
on how to assess this definitively?

Anyway, here's hoping there's some actual data available, and it's a
good one for my needs.

Ed

I fired one up, and found the following info:

Vs +12VDC works just fine, draws about 400 mA cold start, down to 150 mA
warmed up. Initial current draw rises with input voltage, up to about
11V, then constant, viewed up to 16 V.

Cold start frequency less than 9.999953 MHz, reaching normal in about
5-10 minutes.

Output is sine - haven't measured accurately yet but seems about right
on scope for the typical +9 dBm specs listed.

Tune reference about +8 V.

Frequency tune range about +12/-17 Hz, at Vt = 0V/+8V (Vr).

!0 MHz "exact" is at Vt = +3.096 V

Ed

Been busy planning and rounding up parts and subsystems for my
multi-function box for looking at phase noise and stability etc mostly
on 10 and 5 MHz. The PN part will be kind of like Erik's recent project,
using those identical MTI 10 MHz OCXOs (the subject of my recent info
search) for initial experimenting. The baseband analyzer will be the
HP3561A (0-100 kHz).

The other part will be a fairly simple analog DMTD system, with fixed
100 Hz offset for 10 MHz, and 50 Hz for 5 MHz. I found I have a nice old
10 MHz OCXO that has mechanical tuning only, that will be committed
(built in) to this project. It's set now for 9.9999 MHz (10 MHz - 100
Hz). I have no suitable 5 MHz OCXOs for this, so figured on dividing the
9.9999 MHz by two when needed.

I recall many times over the years encountering talk about low noise
regenerative dividers, but now that I'm actually contemplating making
one, I can't seem to find much detail. I found a number of commercial
ones like at Wenzel et al, but not much in the way of design detail.
John M's ke5fx site has a good example for 80-40-20 MHz, but I can't
seem to find any of the linked papers and such that were the basis. What
I'm hoping is to find one or more examples of designs already figured
out for 10 MHz/2. I think there must be some out there.

Does anyone know of such?

BTW I finally pulled the trigger on acquiring an HP8663A, which I've
been wanting to get for a very long time. I've had a fully loaded (all
bands) HP11729C for many years, to go with it, and finally be of use. In
the TIM department, I have a busted HP5370A, and a good HP5372A.

Ed

Ed,
You might look at the paper developed by Luciano Paramithiotti for a 10MHz to 5MHz regenerative divider. He has kindly shared his design, complete with circuit diagram. Might be what you're looking for.
[ http://www.timeok.it/wp-content/uploads/2015/08/low-noise-regenerative-divider1.pdf | http://www.timeok.it/wp-content/uploads/2015/08/low-noise-regenerative-divider1.pdf ]

He has lots of other time/frequency related stuff on his website as well.
http://www.timeok.it/time-and-frequency/

Cheers,
DaveM

From: "Time-Nuts" time-nuts@lists.febo.com
To: "Time-Nuts" time-nuts@lists.febo.com
Cc: "ed breya" eb@telight.com
Sent: Friday, July 29, 2022 8:30:14 PM
Subject: [time-nuts] Looking to build 10 MHz tp 5 MHz regenerative frequency divider

Been busy planning and rounding up parts and subsystems for my
multi-function box for looking at phase noise and stability etc mostly
on 10 and 5 MHz. The PN part will be kind of like Erik's recent project,
using those identical MTI 10 MHz OCXOs (the subject of my recent info
search) for initial experimenting. The baseband analyzer will be the
HP3561A (0-100 kHz).

The other part will be a fairly simple analog DMTD system, with fixed
100 Hz offset for 10 MHz, and 50 Hz for 5 MHz. I found I have a nice old
10 MHz OCXO that has mechanical tuning only, that will be committed
(built in) to this project. It's set now for 9.9999 MHz (10 MHz - 100
Hz). I have no suitable 5 MHz OCXOs for this, so figured on dividing the
9.9999 MHz by two when needed.

I recall many times over the years encountering talk about low noise
regenerative dividers, but now that I'm actually contemplating making
one, I can't seem to find much detail. I found a number of commercial
ones like at Wenzel et al, but not much in the way of design detail.
John M's ke5fx site has a good example for 80-40-20 MHz, but I can't
seem to find any of the linked papers and such that were the basis. What
I'm hoping is to find one or more examples of designs already figured
out for 10 MHz/2. I think there must be some out there.

Does anyone know of such?
Ed

time-nuts mailing list -- time-nuts@lists.febo.com
To unsubscribe send an email to time-nuts-leave@lists.febo.com

Just wondering if anyone happens to know typical or an actual measured
phase noise instance of the 640 MHz reference in the HP8663A. It does
not seem to be specified (unless I missed it somehow), but there's an
"option 03" that specifies it. I don't think the option provides any
improvement, just results of an actual measurement on the particular unit.

Ed

Thanks all, for replies and possibly running some tests. The reason I'm
asking about this is that now that I finally have the HP8663A, I'd like
to rig it up with the HP11179C, and its documentation mentions all this
option 3 stuff on the 8663A. So I started wondering, just how good is
it, and if there's a "typical" or representative spec. Again, I believe
option 3 just means that the particular unit was tested at the factory
before delivery, and the results documented and included with the unit.
Over time, this info could be lost.

My unit has option 2 only, which is fancier phase modulation, that could
be handy for some things. Before it arrived, I studied up the design and
evolution in the manuals. It appears that the 8662A and earlier 8663A
models may have had the 10544 OCXO installed, while mine is one of the
later versions, officially sporting the 10811-60111. I found that this
is indeed what's in there, and the cal seal is marked ca 2014 - likely
the last time it was messed with. It seems to work so far, and it is
quite an impressive beast.

On the 11179C, I had a stroll down memory lane for a while. I kind of
forgot how much I had worked on it about a dozen years ago. Since I had
no prospect of getting a 8662A or 8663A back then, I had been collecting
and mounting parts to make a fairly high grade built-in 640 MHz source.
The scheme was based on a 80 MHz VTOCXO, to be doubled up and filtered
thrice to 640 MHz, and phase locked to 10 MHz if needed. I already had
all the major pieces fitted, and a bunch of notes on the design details.

I started yanking all this stuff out to restore it to stock, but then
started thinking "not so fast - maybe this could be pretty good after
all." Since I really don't know how good the 640 MHz from the 8663A is,
I'll keep all the pieces and info from the other scheme together, just
in case.

Ed

Thank you John Miles for putting together the great summary of 8663A
comparisons and spec info. That pretty much tells the story. I think the
bottom line is that any 8663A will be fairly close (within 10 dB) to the
option 3 description, but no guarantee.

Ed

Tobias wrote

"Hi Ed, could you share some info about your own 640 MHz source?
I would be very interested.
I do have an 8663A, but I have no idea about how good it actually is and
currently, I don't have a possibility to test it because I still lack an
11729C carrier noise test set (but I would like to make something
similar myself some day)."

The plan was to do the doubling and amplifying similar to the
8662A/8663A (which I think are about the same in this respect), except
that I don't have anything like the mentioned 40 MHz and 160 MHz crystal
filters. Starting from the higher OCXO frequency should help some,
depending on its noise performance.

The filters I do have can't fix anything close-in, but should work very
well on spurious content from almost any multiplier scheme. The first
one especially is a single unit 160 MHz BPF made from two tubular
coaxial ones cascaded. I didn't make it - it came this way as a
commercial product. It has enormous stop-band suppression maybe 200 dB
(theoretical) by +/- 10 MHz away from fc, but large insertion loss about
10 dB. The plan was to double the 80 MHz OCXO into this filter, which
should eradicate all spurious from below. The 320 and 640 MHz BPFs are
more conventional.

The 80 MHz OCXO was apparently quite common around 20-30 years ago.
Despite this, I couldn't find any specs twelve years ago, and can't find
any today, even though there seems to be plenty of them still around and
for sale. The ones I have are Vectron 229-9237, and 229-5657-1,
apparently the same except for mechanical construction.

I have a bunch of similar units, mostly oddball frequencies in the 100
MHz range, and I did have to take some apart (soldered shut cans) over
the years to modify for particular projects. I found they all used
half-frequency crystals and built in doublers. The 80 MHz is no
exception - a quick look on the SA shows it's a 40 MHz OCXO that's
doubled up internally, so it's really only starting with a four times
frequency versus multiplying a 10 MHz reference.

Another thing I noticed is that the 640 MHz SAW BPF in the 11729C may
not be for closer in spurious content, but mostly wide cleanup, and
optimized to form a good oscillator when used for that mode. The manual
says that the purpose of the filter is to reduce 120 MHz, 520 MHz, and
760 MHz spurs. These and others naturally come from the rather
complicated 8662A/8663A reference generator/multiplier system, and the
640 MHz output does not appear to have very much isolation from all this
activity.

So, if you make your own 640 MHz "clean and simple" by direct
multiplication, with no side deals for other frequencies, the result
will not include the extra stuff that would be coming from the
generator. If you also start with a good HF OCXO with known specs, and
do careful multiplication, filtering, and PLLing, I think it can beat
the noise performance of the 8662A/8663A's 640 MHz source. How much? I
dunno, but suspect that the hump from around 10 Hz to 10 kHz may be due
in part to all the reference making and synthesizing action going on in
there, that's somehow included in the 640 MHz output. That is, presuming
the 10 MHz internal reference has no such hump. If it does, then it
could be simply the result of the multiplication factor, and unavoidable.

Ed

Going through some parts inventory and sorting exercise, I have a bunch
of these MLP-102 mixers, salvaged from a bunch of nearly identical RF
modules long ago. The environment was around 1 GHz RF and LO or a little
more, and 70 MHz IF. They are in the common 8-pin relay can style like
MCL SRA-1.

I believe that I found some meager data on them many years ago, but
trying recently, nothing whatsoever. Does anyone happen to have such
info, like from an old (1980s) KDI RF parts catalog? Some are marked
EMCO, and some KDI/EMCO.

Ed

Maybe EMCO (sales agent in Germany) can help you.

Here are the contact data:

EMCO Elektronik GmbH
Fraunhoferstrasse 14
D-82152 Planegg/Germany
Tel +49 (0)89 895 5650
Fax +49 (0)89 895 56510
info@emco-elektronik.de

Best regards
Bernd

-----Ursprüngliche Nachricht-----
Von: ed breya via time-nuts [mailto:time-nuts@lists.febo.com]
Gesendet: Donnerstag, 25. August 2022 07:29
An: time-nuts@lists.febo.com
Cc: ed breya eb@telight.com
Betreff: [time-nuts] Looking for info on KDI/EMCO MLP-102 DBM

Going through some parts inventory and sorting exercise, I have a bunch of
these MLP-102 mixers, salvaged from a bunch of nearly identical RF modules
long ago. The environment was around 1 GHz RF and LO or a little more, and
70 MHz IF. They are in the common 8-pin relay can style like MCL SRA-1.

I believe that I found some meager data on them many years ago, but trying
recently, nothing whatsoever. Does anyone happen to have such info, like
from an old (1980s) KDI RF parts catalog? Some are marked EMCO, and some
KDI/EMCO.

Ed

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Since it appears there's no nice and easy way to access the OCXO coarse
tuning cap (there almost certainly is one in there, whether it's
accessible or not), a practical fix is a simple circuit mod (add one
resistor) to the DAC circuit.

I looked up the AD1861, and see that it has the typical built in opamp
and feedback resistor for current to voltage conversion. Also,
apparently, depending on hookup and the data formatting, it can be
arranged to go between +/- 3V output. The simplest way to fix the output
range issue, without changing the basic hookup, any software, or VCO
gain factors, is to just add pure DC offset current at the summing node.

If you need the output voltage to be more plus, then pull some current
from the summing node into the -5V supply through a resistor. The FS
current from the DAC seems to be around 1-2 mA, so a value for R in the
few to tens of k-ohms range can move it significantly. Just figure out
what the scaling actually is, and set it up for enough offset to keep
the range good for a while - you can adjust again as it ages. Two things
that will be different from original is that the external offset R won't
track the internal feedback R thermally, and using the minus 5V for a
reference may not be as clean as you'd want. The data sheet I found
doesn't say much about the internal reference, but I'm guessing it's a
+1.25 or 2.5V bandgap. Or, it may simply be the +5V supply.

Since it's overall in a closed loop feedback system, the offset R
tracking is probably irrelevant, but the noise on the -5V may be an
issue, depending on the PLL characteristics. First prove the concept
with what you already have, then refine it as needed.

Good luck.

Ed